Interface for MEMS intertial sensors

ABSTRACT

In a high-performance interface circuit for micro-electromechanical (MEMS) inertial sensors, an excitation signal (used to detect capacitance variation) is used to control the value of an actuation signal bit stream to allow the dynamic range of both actuation and detection paths to be maximized and to prevent folding of high frequency components of the actuation bit stream due to mixing with the excitation signal. In another aspect, the effects of coupling between actuation signals and detection signals may be overcome by performing a disable/reset of at least one of and preferably both of the detection circuitry and the MEMS detection electrodes during actuation signal transitions. In a still further aspect, to get a demodulated signal to have a low DC component, fine phase adjustment may be achieved by configuring filters within the sense and drive paths to have slightly different center frequencies and hence slightly different delays.

RELATED APPLICATIONS

This application claims benefit of U.S. Application 61/382,898 titledSELF-CLOCKED ASIC INTERFACE FOR MEMS INERTIAL SENSORS filed 14 Sep.2010, incorporated herein by reference.

BACKGROUND OF THE INVENTION

Numerous applications of micro-electromechanical (MEMS) inertial sensorsrequire a high-performance ASIC interface. Existing interface techniquesare not fully satisfactory in various respects.

For example, in feedback (e.g., force-feedback) systems where a driveloop is present, an excitation signal is needed for detection ofcapacitance variations in both a sense loop and a drive loop. Theexcitation signal should not affect the actuation applied, for example,to a proof mass (or proof masses) of the MEMS sensor. However, since theexcitation signal is applied to the proof mass, and since actuationcapacitors share the same proof mass with the detection capacitors,therefore the excitation signal affects the actuation signal content anddynamic range.

Another issue relates to undesired coupling that can occur between theactuation stream of one channel and detection paths of the same channel,or even different channels (e.g., sense mode to sense mode coupling orsense mode to drive mode coupling, etc.). Such coupling can distort thesignal and result in severe degradation in the performance of thedetection front-end circuits. This effect is exaggerated in sense mode,as the combined effect of parasitic capacitance and process mismatch ison the order of the detection capacitance variation.

Several solutions have been proposed to solve this coupling issue. Somesolutions depend on frequency separation between actuation and detection(which works only in the case of coupling between different channels);other solutions depend on estimating the coupling transfer function andcompensating this effect in later stages (in digital domain signalprocessing or—at the cost of increased complexity—in the analog domain).Other proposals have included decreasing the actuation signal level (atthe expense of reduced actuation dynamic range), or manual trimming tocompensate for the mismatches.

In feedback (e.g., force-feedback) systems where a drive loop ispresent, a sense signal may contain a desired sensor input signalAM-modulated at the frequency of a drive signal. Hence, to demodulatethe bit stream to get the original signal, the drive and the sensesignals are multiplied using a multiplier to obtain a demodulated outputsignal. To get the demodulated signal to have the lowest possible DCcomponent, accurate phase adjustment between SNS and DRV bit streams maybe required. Various approaches to achieving this phase adjustmenttypically entail power and/or area penalties.

Hence, an improved interface for interfacing to MEMS inertial sensors isdesired.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

The present invention may be further understood from the followingDetailed Description in conjunction with the appended drawing figures.In the drawing:

FIG. 1 is an architecture block diagram of a circuit (e.g., an ASIC) inwhich the present interface techniques may be used.

FIG. 2A is a diagram of a drive loop of the ASIC of FIG. 1.

FIG. 2B is a diagram of a sense loop of the ASIC of FIG. 1.

FIG. 3 is diagram of a MEMS electrical model together with detectioncircuits of sense loop and drive loop.

FIG. 4 is a signal plot of a sample actuation spectrum showing highfrequency components.

FIG. 5 is a waveform diagram illustrating actuation and detectionwaveforms and the consequent effective actuation force.

FIG. 6 is diagram of a MEMS electrical model illustrating a modifiedactuation technique in which the excitation signal controls the shape ofthe actuation signal.

FIG. 7 is a waveform diagram illustrating modified actuation anddetection waveforms and the consequent effective actuation force inaccordance with an embodiment of FIG. 6.

FIG. 8 is a waveform diagram like that of FIG. 7 showing waveforms inaccordance with a RZ (return-to-zero) option.

FIG. 9 is a diagram illustrating unintended coupling from an actuationpath to a detection path.

FIG. 10 is a waveform diagram showing actuation and excitation signalstogether with an added reset signal RST.

FIG. 11 is a diagram showing the reset signal RST applied to detectioncircuits C/V SNS and C/V DRV and to MEMS electrodes.

FIG. 12 is a waveform diagram illustrating the modified actuation anddetection waveforms of FIG. 7 used in combination with the reset signalof FIG. 10.

FIG. 13 is a waveform diagram like that of FIG. 12 showing waveforms inaccordance with a RZ (return-to-zero) option.

FIG. 14 is a block diagram of a demodulation portion of the circuit ofFIG. 1.

FIG. 15 is a plot showing magnitude and phase responses of the filtersof FIG. 14.

FIG. 16 is a plot of a phase shift as a function of filter centerfrequency for an offset one of the bandpass filters of FIG. 15.

DETAILED DESCRIPTION SUMMARY

This patent disclosure describes an ASIC or other circuit to interfacewith MEMS inertial sensors such as vibratory MEMS gyroscopes andaccelerometers, including in closed loop configurations. Closed loopconfiguration provides best performance in harsh environments.Techniques to improve the sensor interface performance are described,including techniques that allow for extending dynamic range of the MEMSactuation and detection signals, canceling coupling between electrodesin the MEMS module, and achieving fine phase tuning between sense anddrive loops for improved demodulation.

Description

The system architecture of an exemplary ASIC is shown in FIG. 1.Functionally, this system can be divided into main blocks as follows:

1—MEMS Sensor Interface Circuitry (110, 120, 130):

This part of the circuit provides actuation voltages for sense and driveelectrodes of the MEMS sensor, through sense actuation (SNS ACT)switches 121 and drive actuation (DRV ACT) switches 111, respectively.Moreover block 130 (PM EXC) provides a proof mass excitation voltagerequired by capacitance sensing circuits in both the drive and senseloops. Finally, a sense capacitance-to-voltage converter 123 (C/V SNS)and a drive capacitance-to-voltage converter 113 (C/V DRY) performcapacitance-to-voltage sensing in the corresponding loops.

2—MEMS Sensor Drive (DRY) Loop 110:

Referring to FIG. 2A, the drive loop 110 incorporates a phase shift 115A(DRY Phase) required to achieve an oscillation condition of a driveresonator 119 (including a mechanical element, e.g., a suspended mass)of the MEMS sensor, as well as automatic gain control 115B (AGC) tocontrol the amplitude of oscillation of the mechanical element. In theillustrated embodiment, an ADC 117 (which may be implemented, forexample, as a band pass sigma delta modulator) converts the drive loopC/V output from circuit 113 into a single bit reading of the MEMS sensordrive oscillation signal.

3—MEMS sensor sense (SNS) loop 120:

Referring to FIG. 2B, in the illustrated embodiment, the MEMS sensorincludes a sense resonator 129 (including a mechanical element, e.g., asuspended mass). The sense resonator 129 is acted upon by an inputsignal to be sensed (for example, motion) and a feedback signal outputof the sense actuation switches 121. As described more fullyhereinafter, in the illustrated embodiment, an excitation signal EXCoutput by block 130 (FIG. 1) is applied to both the drive resonator 119and the sense resonator 129.

The sense loop 120 performs feedback (e.g., force-feedback) motioncontrol of a mechanical element of the sense resonator 129, and providesa digital output reading. This is achieved by implementing closed-loopfeedback (e.g., force-feedback) using an electro-mechanical sigma deltamodulator 120′ for the sense mode. The ASIC can also be configured tooperate in open loop mode. In this case, the sense loop is opened(switch 129, FIG. 1) and the electronic filter H_(x)(z) coefficients arealtered, such that the electronic filter 125 operates as a sigma-deltaADC.

4—The Digital Processing Core (140, FIG. 1): The digital processing core140 decimates and filters (141) the output of the sigma delta modulatorsof both the drive and sense loops (110, 120), and performs a final sensesignal demodulation operation (143). In addition, the digital processingcore 140 performs temperature compensation (145) of the MEMS sensorreading, and controls an SPI interface 150. In one aspect, the digitalprocessing core 140 functions as an electrostatic actuation controller.Additionally, the digital processing core 140 may output a reset signal,disable signal, or power down signal used to reduce undesired couplingof actuation signals as described below.

5—Power Management (160, FIG. 1):

This block provides all required biasing currents and supply voltages todifferent circuit blocks. Moreover, it generates required high-voltageactuation reference signals. In the illustrated embodiment, a band gapreference voltage Vref (161) is generated and buffered (163, 165) forthe ADCs (117, 125) and for MEMS sensor excitation and actuation (111,121, 130). The power management block 160 is also responsible forgenerating necessary voltages for operation of a ROM 170. In theillustrated embodiment, a charge pump 180 is used for this purpose.

6—The Temperature Sensing System (185, FIG. 1):

This block senses the die temperature and converts it into a digitalreading.

7—Clocking PLL (190, FIG. 1)

The PLL generates the master clock of the system.

ASIC Self-Clocking

In one embodiment, an ASIC self-clocking technique is used. Thistechnique simplifies interfacing of the ASIC to different MEMS sensormodules.

Feedback (e.g., force-feedback) operation reduces system sensitivity toMEMS sensor process variations, increases bandwidth, and allowsoperation in matched mode. Therefore, a closed loop configuration canachieve best performance in harsh environments. Incorporating the MEMSsensor as one part of the feedback (e.g., force-feedback) loop filterconverts the system into a hybrid electromechanical ΣΔ modulator with acontinuous-time (CT) part represented by the mechanical filter and adiscrete-time (DT) part represented by the electronic filter. The CTnature of the mechanical filter makes the performance of the ΣΔmodulator sensitive to the exact feedback pulse shape. For this reason,a low-jitter clock is required for best performance using forcefeedback.

The ASIC of FIG. 1 uses a PLL 190 to generate the system clock. The highQ resonance of the MEMS sensor oscillation (of the drive resonator 119,for example) may be used to generate a clean reference clock for theASIC PLL 190, resulting in a low noise floor at the ASIC output. Hence,the ASIC PLL 190 does not need an external crystal as a reference clock,and the ASIC becomes self-clocked. Using a MEMS sensor based referenceclock allows fixing the ratio between the system sampling frequency andthe mechanical drive resonance frequency (e.g., Coriolis signalcarrier). Therefore, this arrangement enables interfacing to awide-range of MEMS sensor modules. More particularly, the self-clockingtechnique in the ASIC allows the drive loop 110, sense loop 120 and DSPblocks 140 to track the frequency of the MEMS sensor. Hence, the ASICcan interface to a wide range of MEMS sensor frequencies.

The drive loop of FIG. 2A, described above, works in conjunction withthe PLL 190 to enable self-clocking. Since the ASIC is self-clocked, theADC sampling frequency is locked to the MEMS sensor resonance frequencythrough the PLL 190, which controls the sampling frequency of the ADC117.

Low jitter self-clocking of the ASIC may be taken advantage of by theMEMS sensor sense loop. In the illustrated embodiment, the MEMS sensorsense loop of FIG. 2B, described above, is implemented based oncontinuous-time ΔΣ modulation, because the low -jitter clocking of theself-clocked ASIC achieves better SNR for CT feedback (e.g.,force-feedback) operation. As an example, the illustrated sense loop isachieved by implementing a 4th order electro-mechanical ΔΣ modulatorusing a switch-cap electronic filter 125. The modulator architecture maybe based on a feed-forward topology with a feed-back branch to stabilizethe loop. The noise floor appearing at the output is minimized, as aresult of the low -jitter clocking effect on continuous-time ΔΣoperation being minimized. Moreover, the ASIC can interface to a rangeof MEMS sensor modules by using a programmable divider in the PLL 190.

The digital processing core 140 of FIG. 1, described above, filters theoutput of the ΔΣ modulators of both the drive and sense loops (110,120), and performs the final output demodulation using the demodulator143. The demodulation output is decimated using a programmabledecimation filter 141. Two band pass filters (1405, FIG. 14), centeredat the MEMS sensor resonance frequency, filter out noise of both thedrive and sense loops, before multiplying them for demodulation, toavoid mixing and down conversion of quantization noise in the band ofinterest. The poles and zeros of each band pass filter (1405) scale withthe sampling frequency (depending on the MEMS sensor module), andtherefore the center frequency will be correctly tuned with variationsof the MEMS sensor resonance frequency.

Actuation Technique with Improved Dynamic Range

In the illustrated embodiment, the excitation signal EXC (FIG. 1) isapplied to both the drive resonator 119 and the sense resonator 129 andis needed for the detection of capacitance variations in both the senseand drive loops (120, 110). This excitation signal should not reduce theeffectiveness of the actuation signals (SNSACT, DRYACT).

FIG. 3 illustrates, for example, the case of a fully differential MEMSsensor showing an arrangement of the electrodes. In the sense loop,variable capacitances associated with a sense resonator (129, FIG. 2B)are arranged in a “N” configuration. First and second actuationcapacitors Ca1 and Ca2 are arranged in the arms of the Pi; sensecapacitors Cs1 and Cs2 are arranged in the legs of the Pi and arecoupled to inputs of the sense C/V converter 123. The excitation signalEXC (also referred to as a proof mass voltage V_(PM)) is applied betweenthe arms of the Pi. A similar configuration applies to the driveresonator and the drive loop.

Since the excitation signal EXC is added on proof masses of therespective resonators, and since the actuation capacitors (Ca) share thesame proof mass with the detection capacitors (Cs), therefore theexcitation signal affects the actuation signal content and dynamicrange.

In the following electrostatic actuation technique, the excitationsignal (used to detect capacitance variations) is used to control thevalue of the actuation signal bit stream to allow the dynamic range ofboth actuation and detection paths to be maximized and to preventfolding of high frequency components of the actuation bit stream due tomixing with the excitation signal.

In the illustrated embodiment, the electrostatic actuation forcegenerated in response to an excitation signal can be calculatedaccording to the following equation, wherein the excitation signal EXCis represented as Vpm:

F_(act)α((V_(act+)−V_(pm))²−(V_(act−)−V_(pm))²)   (1)

F_(act)α(V_(act+) ¹−V_(act−) ²−2*V_(pm)(V_(act+)−V_(act−)))   (2)

Note that the third term in the (2) will cause both degradation inactuation dynamic range and folding of high order components of theactuation signal due to mixing with the excitation signal. FIG. 4 showsan example of the resulting high order components of a sample actuationstream.

Past systems have usually been designed to compromise between the PMsignal level (V_(pm)) and the actuation voltage level, which will eitherdecrease the detection dynamic range if the V_(pm) value is reduced orthe actuation dynamic range if the V_(pm) value is increased. FIG. 5shows the waveforms of one actuation and detection scheme. In thisexample the actuation signal is assumed to have a voltage level(V_(ref1)), while the excitation signal is assumed to have a voltagelevel (V_(ref2)). If V_(ref1)=V_(ref2), then the effective actuationforce dynamic range is equal to zero (positive and negative streams havethe same waveform).

Assuming that V_(ref1)=V_(ref) and V_(ref2)=Vref/2 as a compromisebetween actuation and detection dynamic ranges, a degradation of 6 dB isintroduced to both the actuation and the detection dynamic ranges.

To avoid such an impairment of dynamic range, the value of the actuationsignal may be controlled according to the current value of the PM signalas shown in FIG. 6. A block 600 performs digital processing andreference generation and may correspond to blocks 140 and 160 of FIG. 1.The block 600 receives sense and drive bit streams (SNS, DRV). The blockalso uses the excitation signal V_(PM) as an input signal to conditionthe actuation signals. In response to the sense and drive bit streams,and taking into account the excitation signal V_(PM), the block 600outputs actuation signals, for example actuation signals Vsact+/Vsact−and Vdact+/Vdaet. The notation Vact+, Vact− is used to refer generallyto these four actuation signals.

For a square wave excitation signal, the actuation signal bit stream maybe logically combined (e.g., XQRed or XNORed) with the PM signal. FIG. 7shows the waveforms of the actuation signals (Vact+, Vact−), excitationsignals and the resulting electrostatic force components (first andsecond terms in equation (1) above). In FIG. 7, actuation bit stream isXORed with a signal PM (V_(pm)) to produce Vact+, while the actuationbit stream is XNORed with the PM signal to produce Vact-. From FIG. 7,it may be seen that no folding of high frequency components occurs(i.e., actuation signal content is preserved intact), and that both theactuation force and excitation signal are represented by the fulldynamic range of the reference voltage.

FIG. 8 shows the actuation and detection waveforms with an addedReturn-to-Zero (RZ) option, which may be used in continuous timeactuation for enhancing the linearity of the DAC. The block 600, in oneaspect, may be considered as circuitry (or means) for applying areturn-to-zero encoding to an actuation signal.

By controlling the actuation stream according to the value of theexcitation signal, the following advantages may be achieved:

-   Allows AC square wave excitation-   Makes the excitation transparent to the actuation-   Eliminates signal Folding-   Maximizes the available actuation dynamic range-   Maximizes the available detection dynamic range-   Allows smaller supply voltages

Actuation Signal Coupling Cancellation

In feedback (e.g., force-feedback) systems, an actuation stream isapplied on the actuation electrodes to produce a certain movement on theproof mass. (This feedback can be in a positive feedback loop or anegative feedback loop). Ideally, the actuation voltage will affect themechanical element according to a response characteristic of themechanical element.

Due to process mismatch and parasitic capacitance, the actuation bitstream can couple directly to the detection circuit as shown in FIG. 9.Unintended coupling paths Ccop1 and Ccop2 couple from the Vact+ signalto the Cs1 and Cs2 signals, respectively. Similarly, unintended couplingpaths Ccom1 and Ccom2 couple from the Vact− signal to the Cs1 and Cs2signals, respectively. This coupling can occur between the actuationstream of one channel and detection paths of the same channel, or evendifferent channels (e.g., SNS mode to SNS mode coupling or SNS mode toDRV mode coupling).

Obviously, these coupling paths can distort the signal and results insevere degradation in the performance of the detection front-endcircuits. This effect is exaggerated in SNS mode, as the parasiticcapacitance/process mismatch is on the order of the detectioncapacitance variation (e.g., Cs1, Cs2 of FIG. 9).

The effects of such coupling may be overcome by performing adisable/reset of at least one of and preferably both of the detectioncircuitry and the MEMS detection electrodes during actuation signaltransitions as shown in FIG. 11, for example using a reset signal RST asshown in FIG. 10. More generally, a control signal may be used to reset,disable, or power down selected circuit elements. In the embodiment ofFIG. 11, a reset signal RST is applied to the C/V SNS detectioncircuitry 123 and to the C/V DRV detection circuitry 113. In addition,analog switches SWs and SWd are provided between the sense electrodes ofthe MEMS sensor and the drive electrodes of the MEMS sensor. The resetsignal RST is used to control the switches SWs and SWd, causing them toclose when the reset signal is asserted. Resetting the detectioncircuitry (113, 123) and the MEMS electrodes cancels the effect of theparasitic coupling paths of the actuation signal.

To allow the detection of the excitation signal edges in the presence ofthe reset pulse RST, a time mismatch (t_(mismatch) FIG. 12) between theexcitation signal and the actuation signal edges may be introduced.Furthermore, the foregoing coupling cancellation technique may be usedtogether with the dynamic range improvement technique describedpreviously. The resulting waveforms are shown in FIG. 12. FIG. 13 showsa sample timing diagram where: 1) a mismatch in timing between theactuation and detection signals is added; 2) XOR actuation with RZ isapplied as previously described; and 3) a reset/disable pulse RST forthe detection electrodes and circuitry is added around the actuationsignal edges for XOR actuation.

Using the foregoing technique to cancel the effect of the parasitic pathbetween actuation and detection in feedback (e.g., force-feedback)systems, the following advantages may be achieved:

-   Allows ΣΔ actuation signal-   Removes coupling between actuation signals and different detection    channels-   Maximizes the available actuation dynamic range-   Avoids need to decrease actuation signal to decrease the coupled    signal-   Maximizes the available detection dynamic range-   Allows detection dynamic range to be completely allocated for    detection signal-   Removes need for complex trimming-   Removes the need for complex signal processing in analog or digital    domains

Demodulation with fine phase tuning between sense and drive loops

Referring to FIG. 14 and FIG. 2B, the SNS sigma delta bit stream 1401 atthe SNS loop output (out, FIG. 2B) contains the input signal (inputsignal, FIG. 2B) AM-modulated at the DRV signal frequency. Hence, todemodulate the bit stream to get the original signal, the DRY and theSNS signals (1401, 1402) are multiplied using a multiplier 1403 toobtain a signal Demodulation Out as shown in FIG. 14. Two band passfilters (BPF) 1405 s and 1405 d may be used to remove sigma delta noisebefore multiplication. To get the demodulated signal to have the lowestpossible DC component, phase adjustment (1407, FIG. 14) between the SNSand DRV bit streams (1401, 1402) may be applied. This phase adjustmentcan be achieved by adding delay to either the SNS bit stream 1401 or theDRY bit stream 1402. In one embodiment, the phase adjustment 1407 may beimplemented using a programmable shift register—i.e., the signal can bedelayed by multiples of the operating clock periods. However, thistechnique can add only coarse phase shift if the system clock period isrelatively large. For example, if the MEMS sensor resonance is at 3 kHzand the system clock frequency is 400 kHz, then the minimum phase shiftthat can be added using such a shift register is:

${Phase}_{minimum} = {\frac{360}{400\mspace{14mu} {{kHz}/3}{kHz}} = 2.7^{o}}$

If the required phase shift is about 87°, then the shift register lengthshould be greater than 32 bits. In order to get fine tuning in the phaseadjustment, the clock frequency can be increased; however, thisincreases the power consumption of the system and the design complexitywith respect to timing. Moreover, the area increases as a result ofadding registers in the shift register to get the required phase shift.For example, if the required phase resolution is 2.7°/4, then the clockfrequency should increase 4 times. In addition, if the required phaseshift is about 87°, then the shift register length should be greaterthan 128 bits, which is four times the length of the shift register inthe previous case.

In order to obtain accurate phase alignment without incurringsignificant power or area penalty, the BPFs 1405 (that are already usedto remove the SD noise) may be configured to achieve fine phase tuningby introducing a certain offset in center frequencies of the BPFs. Theshift register 1407 may be kept for coarse tuning.

As an example, the transfer function of filters that may be used in theASIC of FIG. 1 is shown in FIG. 15. In the illustrated embodiment, bothfilters are identical with bandwidths of 200 Hz centered at the MEMSsensor resonance. Hence, the phase added by the respective filters tothe SNS and the DRY signals is exactly the same phase. To add finetuning in the phase adjustment, the center frequencies of the twofilters can be made different from each other with a very smalldifference. As shown in FIG. 15 the phase response of the filter is verysharp at the center frequency. As a result, shifting the center of thedrive (or sense) filter slightly can add the required phase shift. Forexample, changing the center frequency of one of the filters slightlymight mean that one of them can be centered at 3 kHz, whereas the otheris centered at 3.01 kHz. Therefore, the two filters coincideapproximately with respect to magnitude, whereas the phase shifts of thetwo filters will be different so as to result in a net phase shift ofthe required value.

As shown in FIG. 16, if the center frequencies of the filters differ by10 Hz, then the phase shift between the two filters is about 0.04 rad(˜2.7°). In the illustrated embodiment, 2.7° is the maximum phase shiftrequired to be added by the filter, since phase shift in increments of2.7° can be added directly by coarse tuning (using, for example, a shiftregister as described). As a rough estimate therefore, in theillustrated embodiment, the center frequencies of the two filters maydiffer by less than 10 Hz to get a fine phase shift less than 2.7°.

In some embodiments, the center frequencies may be fixed. In otherembodiments, the center frequencies may be adjustable.

Using the foregoing technique to achieve fine phase adjustment betweensense and drive paths in feedback (e.g., force-feedback) systems, thefollowing advantages may be achieved:

-   Allows a demodulated output with a low DC component to be obtained-   Avoids power and area penalties imposed by other approaches

It will be apparent to those of ordinary skill in the art that thepresent invention can be embodied in other specific forms withoutdeparting from the spirit or essential character thereof The foregoingdescription is therefore to be regarded as illustrative, notrestrictive. The scope of the invention is defined by the appendedclaims, not the foregoing description, and all changes which some withinthe range of scope of equivalents thereof are intended to be embracedtherein.

1. An integrated circuit for interfacing with a MEMS sensor, comprising:an electrostatic actuation controller configured to generate anactuation signal and an excitation signal and to receive a senseinformation signal; and a sensing capacitance-to-voltage converterresponsive to a signal from the MEMS sensor for producing a sensesignal; wherein the electrostatic actuation controller is configured to,during at least some signal transitions of the actuation signal, apply acontrol signal to the sensing capacitance-to-voltage converter to causeit to be reset, disabled or powered down.
 2. The apparatus of claim 1,comprising: a drive capacitance-to-voltage converter responsive to asignal from the MEMS sensor for producing a drive signal; and ademodulator coupled to the drive signal and the sense signal fordemodulating sense information carried by the sense signal to producethe sense information signal.
 3. The apparatus of claim 1, wherein theelectrostatic actuation controller is configured for timing theexcitation signal and the actuation signal such that transitions of theactuation signal occur slightly prior to transitions of the excitationsignal by a time t_(mismatch).
 4. The apparatus of claim 1, wherein theelectrostatic actuation controller is configured for timing theexcitation signal and the control signal such that the excitation signalis caused to transition in response to a transition of the controlsignal to an inactive state.
 5. A method of interfacing with a MEMSsensor comprising a suspended mass, the method comprising: responsive toa sense signal output by the MEMS sensor, sensing capacitance variationscaused by motion of the mass; producing an actuation signal to beapplied to the MEMS sensor for causing an influence to be exerted uponthe mass, and an excitation signal to be applied to at least one sensecapacitor of the MEMS sensor; and during at least some signaltransitions of the actuation signal, resetting, disabling or poweringdown said sensing in response to a control signal.
 6. The method ofclaim 5, comprising timing the excitation signal and the actuationsignal such that transitions of the actuation signal occur slightlyprior to transitions of the excitation signal by a time t_(mismatch). 7.The method of claim 5, comprising timing the excitation signal and thecontrol signal such that the excitation signal is caused to transitionin response to a transition of the control signal to an inactive state.8. The method of claim 5, comprising applying a logical operation to theexcitation signal and a further signal to obtain the actuation signal.9. The method of claim 8, wherein the logical operation is one of an XORand an XNOR operation.
 10. The method of claim 9, comprising applying areturn-to-zero encoding to the actuation signal.
 11. The method of claim5, wherein sensing capacitance variations is performed using acapacitance-to-voltage converter, comprising resetting thecapacitance-to-voltage converter in response to the control signal. 12.The method of claim 11, comprising setting the sense signal to knownvalue in response to the control signal.
 13. The method of claim 5,comprising setting the sense signal to known value in response to thecontrol signal.
 14. The method of claim 13, wherein said sensingcapacitance variations is performed using a capacitance-to-voltageconverter, comprising resetting the capacitance-to-voltage converter inresponse to the control signal.
 15. The method of claim 13, wherein thesense signal is a differential signal comprising a positive signal and anegative signal, the method comprising setting the positive signal andthe negative signal to be equal.
 16. An integrated circuit forinterfacing with a MEMS sensor comprising a suspended mass, theintegrated circuit comprising: sensing circuitry responsive to a sensesignal output by the MEMS sensor for sensing capacitance variationscaused by motion of the mass; control circuitry for applying to the MEMSsensor an actuation signal to cause an influence to be exerted upon themass, and an excitation signal to be applied to at least one sensecapacitor of the MEMS sensor; and circuitry for, during at least somesignal transitions of the actuation signal, applying a control signal tothe sensing circuitry to reset, disable or power down said sensingcircuitry.
 17. The apparatus of claim 16, wherein the control circuitryis configured for timing the excitation signal and the actuation signalsuch that transitions of the actuation signal occur slightly prior totransitions of the excitation signal by a time t_(mismatch).
 18. Theapparatus of claim 16, wherein the control circuitry and the resetcircuit are configured for timing the excitation signal and the controlsignal such that the excitation signal is caused to transition inresponse to a transition of the control signal to an inactive state. 19.The apparatus of claim 16, comprising circuitry for applying a logicaloperation to the excitation signal and a further signal to obtain theactuation signal.
 20. The apparatus of claim 19, wherein the logicaloperation is one of an XOR and an XNOR operation.
 21. The apparatus ofclaim 20, comprising circuitry for applying a return-to-zero encoding tothe actuation signal.
 22. The apparatus of claim 16, wherein the sensingcircuitry comprises a capacitance-to-voltage converter, wherein thecapacitance-to-voltage converter is reset in response to the controlsignal.
 23. The apparatus of claim 22, wherein the sensing circuitrycomprises means for setting the sense signal to a known value inresponse to the control signal.
 24. The apparatus of claim 16, whereinthe sensing circuitry comprises means for setting the sense signal to aknown value in response to the control signal.
 25. The apparatus ofclaim 24, wherein said sensing circuitry comprises acapacitance-to-voltage converter, wherein the capacitance-to-voltageconverter is reset in response to the control signal.
 26. The apparatusof claim 24, wherein the sense signal is a differential signalcomprising a positive signal and a negative signal, wherein said meansfor setting is configured to set the positive signal and the negativesignal to be equal.
 27. The apparatus of claim 26, wherein said meansfor setting comprises an analog switch.